Three-phase llc power supply circuit for high voltage bus input

ABSTRACT

A three-phase power supply circuit is provided. The power supply circuit includes three LLC resonant voltage convertors, three step-down transformers, and a bridge rectifier. Each step-down transformer includes a primary and secondary coil, and each primary and secondary coil has a first node and a second node. Each step-down transformer is electrically coupled with one of the three LLC resonant voltage convertors by the first and second nodes of the primary coils. The bridge rectifier is electrically coupled with the first node of the secondary coil of each of the three step-down transformers. The second nodes of the secondary coils of each of the three step-down transformers are electrically coupled together.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation to U.S. application Ser. No.17/105,746 filed Nov. 27, 2020. The entire disclosure of the aboveapplication is incorporated herein by reference.

TECHNICAL FIELD

Aspects of the disclosure are related to electronic components and inparticular to components for three-phase power systems.

TECHNICAL BACKGROUND

Three-phase LLC power converters are commonly used in a variety ofsystems including telecom systems, fast chargers for electric vehicles,and other applications requiring high power density and high efficiency.

These three-phase LLC power converters typically include aninductor/transformer pair for each of the three phases. Since thesecomponents must withstand large switching currents and voltage stresses,they are commonly among the largest components and most expensive withinthe power converter.

Overview

In an embodiment, a three-phase power supply circuit is provided. Thepower supply circuit includes three LLC resonant voltage convertors,three step-down transformers, and a bridge rectifier. Each step-downtransformer includes a primary and secondary coil, and each primary andsecondary coil has a first node and a second node. Each step-downtransformer is electrically coupled with one of the three LLC resonantvoltage convertors by the first and second nodes of the primary coils.The bridge rectifier is electrically coupled with the first node of thesecondary coil of each of the three step-down transformers. The secondnodes of the secondary coils of each of the three step-down transformersare electrically coupled together.

BRIEF DESCRIPTION OF THE DRAWINGS

Many aspects of the disclosure can be better understood with referenceto the following drawings. While several implementations are describedin connection with these drawings, the disclosure is not limited to theimplementations disclosed herein. On the contrary, the intent is tocover all alternatives, modifications, and equivalents.

FIG. 1 illustrates an exemplary single-phase LLC resonant voltageconvertor.

FIG. 2 illustrates an exemplary control circuit of the single-phase LLCresonant voltage convertor of FIG. 1 .

FIG. 3 illustrates exemplary non-overlapping square wave outputs fromthe control circuit of FIG. 2 .

FIG. 4 illustrates exemplary voltages and currents within components ofthe single-phase LLC resonant voltage convertor of FIG. 1 whencontrolled by the control circuit of FIG. 2 .

FIG. 5 illustrates the exemplary non-overlapping square wave outputsfrom the control circuit of FIG. 2 .

FIG. 6 illustrates an exemplary single-phase LLC resonant voltageconvertor.

FIG. 7 illustrates an exemplary three-phase power supply circuitincluding three LLC resonant voltage convertors.

FIG. 8 illustrates an exemplary control circuit for the three-phasepower supply circuit of FIG. 7 .

FIG. 9 illustrates exemplary non-overlapping square wave outputs fromthe control circuit of FIG. 8 .

FIG. 10 illustrates exemplary currents within primary coils of the threetransformers within the three-phase power supply circuit of FIG. 7 .

FIG. 11 illustrates exemplary currents within active switches within thethree LLC resonant voltage convertors within the three-phase powersupply circuit of FIG. 7 .

FIG. 12 illustrates exemplary phase-to-phase and phase-to-star pointvoltages within the secondary coils of the three transformers within thethree-phase power supply circuit of FIG. 7 .

FIG. 13 illustrates an exemplary three-phase power supply circuitincluding an output filter capacitor.

FIG. 14 illustrates an exemplary synchronous bridge rectifier includingsix switches.

FIG. 15 illustrates an exemplary single-phase LLC resonant voltageconvertor for use within the three-phase power supply circuit of FIG. 13.

FIG. 16 illustrates an exemplary three-phase power supply circuit withstep-down transformers external to the three single-phase LLC resonantvoltage convertors.

FIGS. 17A and 17B illustrate an exemplary unified core body for athree-phase power supply circuit such as that of FIG. 16 .

FIGS. 18A and 18B illustrate an exemplary three-phase magnetics assemblyincluding three step-down transformers and three inductors for thethree-phase power supply circuit of FIG. 16 .

FIG. 19 illustrates exemplary magnetic fluxes within each of thetransformers and inductors and a common return leg within an exemplarypower supply circuit incorporating the three-phase magnetics assembly ofFIGS. 18A and 18B.

FIG. 20 illustrates a portion of a three-phase power supply circuitincluding an output inductor.

DETAILED DESCRIPTION

The example embodiments described herein illustrate different methodsfor constructing a three-phase power supply circuit including three LLCresonant voltage convertors suitable for high voltage direct current(DC) inputs. Each LLC resonant voltage convertor is coupled with astep-down transformer, and the secondary coils of the three step-downtransformers are electrically coupled together in a star formation. Thisconfiguration reduces switching current, voltage stress, and transformerprimary root mean square (RMS) currents, resulting in improvedefficiency.

FIG. 1 illustrates an exemplary single-phase LLC resonant voltageconvertor 100. In this example embodiment, input voltage VIN 102 isapplied to inputs of the voltage convertor across capacitors C6 116 andC7 117 which act to divide the input voltage VIN 102 in half since thevalues of C6 116 and C7 117 are the same. In some example embodimentsinput voltage VIN 102 may be provided by a power factor correctioncircuit.

Switch Q1 141 and diode D4 134 make up a first half-bridge across C7 117within the voltage convertor and switch Q2 142 and diode D3 133 make upa second half-bridge across C6 116 within the voltage convertor. DiodesD3 133 and D4 134 are blocking diodes which block current when switchesQ1 141 and Q2 142 are turned on simultaneously. Switch Q1 141 is drivenby isolated driver E7 122 and resistors R29 151 and R32 152. Switch Q2142 is driven by isolated driver E9 123 and resistors R34 153 and R38154. Isolated drivers E7 122 and E9 123 are both driven by square waveAA 106 which is generated by a control circuit illustrated in FIG. 2 anddescribed in detail below.

The maximum voltage stress on switches Q1 141 and Q2 142 is equal tohalf of the input voltage VIN 102, or the voltage across capacitors C6116 and C7 177 respectively, while switch Q5 143 experiences the entirevoltage stress of input voltage VIN 102. In an example embodiment, whenthe input voltage VIN 102 is 440 volts as illustrated here, switches Q1141 and Q2 142 may be rated for 300-400 volts, while Q5 143 is rated for600-650 volts.

Switch Q5 143 is configured to short diodes D3 133 and D4 134 when it isactivated by isolated driver E1 121 and resistors R12 155 and R13 156.Isolated driver E1 121 is driven by square wave BB 108 which isgenerated by a control circuit illustrated in FIG. 2 and described indetail below.

Each half-bridge drives one node of the primary P1 coil of step-downtransformer TX1 146 through a capacitor/inductor pair. The firsthalf-bridge comprising switch Q1 141 and diode D4 134 drives a firstnode of the primary P1 coil of step-down transformer TX1 146 throughsplit resonant components capacitor C2 112 and inductor L1 144,electrically coupled in series. The second half-bridge comprising switchQ2 142 and diode D3 133 drives a second node of the primary P1 coil ofstep-down transformer TX1 146 through split resonant componentscapacitor C1 111 and inductor L5 145, electrically coupled in series.

In other example embodiments, a single resonant tank comprising C1 111and L5 145 may be used, in which case the value of C1 111 will be halfof the value required by the split arrangement illustrated in FIG. 1 ,and the value of L5 145 will be double the value required by the splitarrangement illustrated in FIG. 1 .

In this example embodiment, the secondary S1 coil of step-downtransformer TX1 146 drives a bridge rectifier comprising diodes D1 131,D2 132, D9 135, and D10 136. The output of the bridge rectifier producesoutput voltage VOUT 104 across output filter capacitor C10 113 drivingload resistance RLOAD 157.

In other example embodiments, synchronous rectifiers may be used inplace of diodes D1 131, D2 132, D9 135, and D10 136. This embodiment isillustrated in FIG. 14 and described in detail below.

FIG. 2 illustrates an exemplary control circuit 200 of the single-phaseLLC resonant voltage convertor 100 of FIG. 1 . In this exampleembodiment, control circuit 200 comprises voltage-controlled oscillator204, D flip-flop 212, delay circuits 214 and 216, and two-input ANDgates U1 222, and U2 224.

VCO 204 receives a voltage input 202 from a compensator circuit output(not illustrated) and outputs clock 206 to the CLK input of D flip-flop212. D flip-flop 212 acts to divide the frequency of clock 206 from VCO204 in half

Output Q 208 of D flip-flop 212 drives an input of first delay circuit216 and a first input of first AND gate U2 224. The output 220 of firstdelay circuit 216 drives the second input of first AND gate U2 224.Inverted output QN 210 of D flip-flop 212 drives the D input of Dflip-flop 212 along with an input of second delay circuit 214 and afirst input of second AND gate U1 222. The output 218 of second delaycircuit 214 drives the second input of second AND gate U1 222.

First AND gate U2 224 provides control signal AA 228, and second ANDgate U1 222 provides control signal BB 226. These control signals areprovided to the LLC resonant voltage convertor 100 of FIG. 1 and areconfigured to control the three switches, Q1 141, Q2 142, and Q5 143,within LLC resonant voltage convertor 100. In this example embodiment,control signal AA 228 drives isolated drivers E7 122 and E9 123 which inturn control switches Q1 141 and Q2 142 respectively. Control signal BB226 drives isolated driver E1 121 which in turn controls switch Q5 142.

FIG. 3 illustrates exemplary non-overlapping square wave outputs 300from the control circuit 200 of FIG. 2 . This timing diagram illustratesoutputs AA 320 from first AND gate U2 224, and BB 310 from second ANDgate U1 222 from FIG. 2 . Note that the voltages and times illustratedhere are exemplary, and various embodiments of the present invention mayprovide square wave signals of various amplitudes and frequencies allwithin the scope of the present invention.

FIG. 4 illustrates exemplary voltages and currents within components ofthe single-phase LLC resonant voltage convertor 100 of FIG. 1 whencontrolled by the control circuit 200 of FIG. 2 .

In this example embodiment, waveforms for the drain voltage 402 of Q5143, current 404 of Q5 143, drain voltage 406 of Q1 141 and Q2 142,drain current 408 of Q1 141 and Q2 142, gate voltage 410 of Q5 143, andgate voltage 412 of Q1 141 and Q2 142 are illustrated.

Note that the voltages, currents, and times illustrated here areexemplary, and various embodiments of the present invention may producewaveforms of various amplitudes and frequencies all within the scope ofthe present invention.

FIG. 5 illustrates the exemplary non-overlapping square wave outputs AA228 and BB 226 from the control circuit 200 of FIG. 2 .

In operation, initially AA 228 is high Q1 141 and Q2 142 are conductingand delivering power to the output. At an exemplary time of T0 530, bothQ1 141 and Q2 142 are turned off If the switching frequency is less thanthe resonant frequency, the current through switches Q1 141 and Q2 142will be equal to the magnetizing current (Imag) of the transformer TX1146.

When switches Q1 141 and Q2 142 are turned off, Imag charges the outputcapacitances of Q1 141 and Q2 142 while the output capacitance (Coss) ofQ5 143, and diodes D3 133 and D4 134, will discharge. Once the outputcapacitance (Coss) of Q5 143 is completely discharged its body diodeturns on and the magnetizing current flows through this diode. If Q5 143is turned on at this time, zero-voltage switching (ZVS) may be achieved.

If the switching frequency is higher than the resonant frequency, thenthe current through the switches will be higher than Imag depending onthe load and the charge and discharge will be faster allowing smallerdead time to achieve ZVS.

At time T1 531, Q5 143 is turned on when its body diode is inconduction. C1 111, C2 112, L1 144, and L5 145 resonate, deliveringpower to the primary coil P1 of transformer TX1 146.

At time T2 532, Q5 143 is turned off. As before, Imag charges the outputcapacitance of Q5 143 and the junction capacitance of diodes D3 144 andD4 134 while discharging the output capacitances (Coss) of Q1 141 and Q2142. Once switch Q5 143 is fully charged Imag flows through switches Q1141 and Q2 142.

At time T3 533, switches Q1 141 and Q2 142 are turned on while theirbody diodes are conducting, achieving zero-voltage switching (ZVS).

This cycle repeats at times T4 534 and T5 535 corresponding to times T0530 and T1 531 respectively.

Note that the voltages and times illustrated here are exemplary, andvarious embodiments of the present invention may provide square wavecontrol signals of various amplitudes and frequencies all within thescope of the present invention.

FIG. 6 illustrates an exemplary single-phase LLC resonant voltageconvertor 600. This three-switch single-phase LLC resonant voltageconvertor 600 is very similar to the voltage convertor 100 of FIG. 1 .

In this example embodiment, an input voltage is applied to inputs DC+606 and DC− 608 of the voltage convertor 600. In some exampleembodiments the input voltage may be provided by a power factorcorrection circuit.

Switch Q1 641 and diode D1 631 make up a first half-bridge and switch Q2642 and diode D2 632 make up a second half-bridge. Diodes D1 631 and D2632 are blocking diodes which block current when switches Q1 641 and Q2642 are turned on simultaneously. Resistor R10 657 to ground is includedbetween diodes D1 631 and D2 632.

Switch Q1 141 is driven by isolated driver E1 621 and resistors R1 651and R2 652. Switch Q2 642 is driven by isolated driver E3 622 andresistors R3 653 and R4 654. Isolated drivers E1 621 and E2 622 are bothdriven by square wave AA 602 which is generated by a control circuitillustrated in FIG. 8 and described in detail below.

The maximum voltage stress on switches Q1 641 and Q2 642 is equal tohalf of the input voltage between DC+ 606 and DC− 608, while switch Q3643 experiences the entire voltage stress of the input voltage betweenDC+ 606 and DC− 608. In an example embodiment, when the input voltagebetween DC+ 606 and DC− 608 is 440 volts, switches Q1 641 and Q2 642 maybe rated for 300-400 volts, while Q3 643 is rated for 600-650 volts.

Switch Q3 643 is configured to short diodes D1 631 and D2 632 when it isactivated by isolated driver E3 623 and resistors R5 655 and R6 656.Isolated driver E3 623 is driven by square wave BB 604 which isgenerated by a control circuit illustrated in FIG. 8 and described indetail below.

Each half-bridge drives one node of the primary P1 coil of step-downtransformer TX1 646 through a capacitor/inductor pair. The firsthalf-bridge comprising switch Q1 641 and diode D1 631 drives a firstnode of the primary P1 coil of step-down transformer TX1 646 throughsplit resonant components capacitor C1 661 and inductor L1 644,electrically coupled in series. The second half-bridge comprising switchQ2 642 and diode D2 632 drives a second node of the primary P1 coil ofstep-down transformer TX1 646 through split resonant componentscapacitor C2 662 and inductor L2 645, electrically coupled in series.

Output voltages OUT+ 610 and OUT− 612 are provided on first and secondnodes of the secondary coil S1 of step-down transformer TX1 646. Whilethis example embodiment includes step-down transformer TX1 646 withinLLC resonant voltage convertor 600, other embodiments may providestep-down transformer TX1 646 external to LLC resonant voltage convertor600 as is illustrated by FIG. 15 and described in detail below.

In other example embodiments, a single resonant tank comprising C2 662and L2 645 may be used, in which case the value of C2 662 will be halfof the value required by the split arrangement illustrated in FIG. 6 ,and the value of L2 645 will be double the value required by the splitarrangement illustrated in FIG. 6 .

FIG. 7 illustrates an exemplary three-phase power supply circuit 700including three LLC resonant voltage convertors 600 from FIG. 6 . Thisexample power supply circuit 700 comprises three LLC resonant voltageconvertors 711, 712, and 713, such as the LLC resonant voltage convertor600 of FIG. 6 .

Input voltage VIN is applied to the DC+704 and DC-706 inputs of each ofthe three LLC resonant voltage convertors 711, 712, and 713. VIN isapplied to the DC+ and DC− inputs of the voltage convertors 711, 712,and 713 across capacitors C1 721 and C2 722, which act to divide theinput voltage VIN in half since the values of C1 721 and C2 722 are thesame.

Control signals AA0 760 and BB0 770 are provided to Phase 0 LLC voltageconvertor 711. Control signals AA1 761 and BB1 771 are provided to Phase1 LLC voltage convertor 712. Control signals AA2 762 and BB2 772 areprovided to Phase 2 LLC voltage convertor 713. These six control signalsare generated by a control circuit illustrated in FIG. 8 and describedin detail below.

The OUT+ outputs (from a first node of the secondary coils of thestep-down transformers) of voltage convertors 711, 712, and 713 areprovided to bridge rectifier 702, while the OUT− outputs (from a secondnode of the secondary coils of the step-down transformers) of voltageconvertors 711, 712, and 713 are electrically coupled together in a starformation. In this example embodiment, the OUT+ outputs of voltageconvertors 711, 712, and 713 are the dot endings (or start of thesecondary windings) of the step-down transformers, while the OUT−outputs of voltage convertors 711, 712, and 713 are the finish endingsof the step-down transformers.

Since the secondary coils of the three step-down transformers areconnected in a star formation, the phase-to-phase voltage is double thatof one phase with respect to the star point. Thus, the transformerprimary to secondary ratio needed at the resonant frequency for the sameoutput voltage is double that of the single-phase LLC.

The start endings of the step-down transformers are connected to themidpoint of each leg of the three-phase diode bridge rectifier 702.

In this example embodiment, bridge rectifier 702 comprises diodes D7731, D8 732, D9 733, D10 734, D11 735, and D12 736. In other exampleembodiments, synchronous rectifiers may be used in place of diodes D7731, D8 732, D9 733, D10 734, D11 735, and D12 736. This embodiment isillustrated in FIG. 14 and described in detail below.

The output of bridge rectifier 702 produces output voltage VOUT 708across output filter capacitor C10 741 driving load resistance RLOAD751.

FIG. 8 illustrates an exemplary control circuit 800 for the three-phasepower supply circuit 700 of FIG. 7 . In this example embodiment, a clocksignal 802 from a voltage-controlled oscillator (VCO) is provided to theCLK inputs of three D flip-flops 810, 812, and 814 which are configuredto divide the frequency of the clock signal 802 by three and providethree pulse signals, each offset from each other in phase by 120degrees. Pulse A 804 is provided by the Q output of D flip-flop 810,Pulse B 806 is provided by the inverted QN output of D flip-flop 821,and Pulse C 808 is provided by the Q output of D flip-flop 814.

The Q output of D flip-flop 810 is provided to the D input of Dflip-flop 812, the Q output of D flip-flop 812 is provided to the Dinput of D flip-flop 814, and the inverted QN output of D flip-flop 814is provided to the D input of D flip-flop 810. These three flip-flopsthus act to provide the three pulse signals 804, 806, and 808 with120-degree phase offsets and a frequency equal to ⅓ f the clock signal802 received from the VCO.

Pulse A 804 is provided to the CLK input of D flip-flop 816. D flip-flop816 acts to divide the frequency of Pulse A 804 in half. Output Q 822 ofD flip-flop 816 drives an input of phase 0 first delay circuit 852 and afirst input of phase 0 first AND gate U2 862. The output 836 of phase 0first delay circuit 852 drives the second input of phase 0 first ANDgate U2 862. Inverted output QN 824 of D flip-flop 816 drives the Dinput of D flip-flop 816 along with an input of phase 0 second delaycircuit 851 and a first input of phase 0 second AND gate U1 861. Theoutput 834 of phase 0 second delay circuit 851 drives the second inputof phase 0 second AND gate U1 861.

The output of phase 0 first AND gate U2 862 provides control signal AA0760 to the Phase 0 LLC voltage convertor 711 of FIG. 7 across resistorR2 872. The output of phase 0 second AND gate U1 861 provides controlsignal BBO 770 to the Phase 0 LLC voltage convertor 711 of FIG. 7 acrossresistor R1 871.

Pulse B 806 is provided to the CLK input of D flip-flop 818. D flip-flop818 acts to divide the frequency of Pulse B 806 in half. Output Q 826 ofD flip-flop 818 drives an input of phase 1 first delay circuit 854 and afirst input of phase 1 first AND gate U4 864. The output 840 of phase 1first delay circuit 854 drives the second input of phase 1 first ANDgate U4 864. Inverted output QN 828 of D flip-flop 818 drives the Dinput of D flip-flop 818 along with an input of phase 1 second delaycircuit 853 and a first input of phase 1 second AND gate U3 863. Theoutput 838 of phase 1 second delay circuit 853 drives the second inputof phase 1 second AND gate U3 863.

The output of phase 1 first AND gate U4 864 provides control signal AA1761 to the Phase 1 LLC voltage convertor 712 of FIG. 7 across resistorR4 874. The output of phase 1 second AND gate U3 863 provides controlsignal BB1 771 to the Phase 1 LLC voltage convertor 712 of FIG. 7 acrossresistor R3 873.

Pulse C 808 is provided to the CLK input of D flip-flop 820. D flip-flop820 acts to divide the frequency of Pulse C 808 in half. Output Q 830 ofD flip-flop 820 drives an input of phase 2 first delay circuit 856 and afirst input of phase 2 first AND gate U6 866. The output 844 of phase 2first delay circuit 856 drives the second input of phase 2 first ANDgate U6 866. Inverted output QN 832 of D flip-flop 820 drives the Dinput of D flip-flop 820 along with an input of phase 2 second delaycircuit 855 and a first input of phase 2 second AND gate U5 865. Theoutput 842 of phase 2 second delay circuit 855 drives the second inputof phase 2 second AND gate U5 865.

The output of phase 2 first AND gate U6 866 provides control signal AA2762 to the Phase 2 LLC voltage convertor 713 of FIG. 7 across resistorR6 876. The output of phase 2 second AND gate U5 865 provides controlsignal BB2 772 to the Phase 2 LLC voltage convertor 713 of FIG. 7 acrossresistor R5 875.

FIG. 9 illustrates exemplary non-overlapping square wave outputs fromthe control circuit 800 of FIG. 8 . This timing diagram illustratesoutputs AA0 760 from phase 0 first AND gate U2 862, BB0 770 from phase 0second AND gate U1 861, AA1 761 from phase 1 first AND gate U4 864, BB1771 from phase 1 second AND gate U3 863, AA2 762 from phase 2 first ANDgate U6 866, and BB3 772 from phase 2 second AND gate U5 865 from FIG. 8.

Note that the voltages and times illustrated here are exemplary, andvarious embodiments of the present invention may provide square wavecontrol signals of various amplitudes and frequencies all within thescope of the present invention.

FIG. 10 illustrates exemplary currents within primary coils of the threetransformers within the three-phase power supply circuit 700 of FIG. 7 .

Waveform 1002 illustrates the current through the primary coil oftransformer TX1 within phase 0 LLC voltage convertor 711. Waveform 1004illustrates the current through the primary coil of transformer TX1within phase 1 LLC voltage convertor 712. Waveform 1006 illustrates thecurrent through the primary coil of transformer TX1 within phase 2 LLCvoltage convertor 713.

Note that the currents and times illustrated here are exemplary, andvarious embodiments of the present invention may produce waveforms ofvarious amplitudes and frequencies all within the scope of the presentinvention.

FIG. 11 illustrates exemplary currents within active switches within thethree LLC resonant voltage convertors 711, 712, and 713 within thethree-phase power supply circuit 700 of FIG. 7 .

Waveform 1102 illustrates the current through Q1 641 within phase 0 LLCvoltage convertor 711. Waveform 1104 illustrates the current through Q3643 within phase 0 LLC voltage convertor 711.

Waveform 1106 illustrates the current through Q1 641 within phase 1 LLCvoltage convertor 712. Waveform 1108 illustrates the current through Q3643 within phase 1 LLC voltage convertor 712.

Waveform 1110 illustrates the current through Q1 641 within phase 2 LLCvoltage convertor 713. Waveform 1112 illustrates the current through Q3643 within phase 2 LLC voltage convertor 713.

Note that the currents and times illustrated here are exemplary, andvarious embodiments of the present invention may produce waveforms ofvarious amplitudes and frequencies all within the scope of the presentinvention.

FIG. 12 illustrates exemplary phase-to-phase and phase-to-star pointvoltages within the secondary coils of the three transformers within thethree-phase power supply circuit 700 of FIG. 7 .

Waveform 1202 illustrates the voltage difference between the OUT+signals of two of the LLC voltage convertors of FIG. 7 . Waveform 1204illustrates the voltage difference between the OUT+ signal and theOUT-signal (connected in a star formation) of one of the LLC voltageconvertors of FIG. 7 . Note that the phase-to-phase peak voltage ofwaveform 1202 is equal to the phase-to-star point voltage of waveform1204.

Note that the voltages and times illustrated here are exemplary, andvarious embodiments of the present invention may produce waveforms ofvarious amplitudes and frequencies all within the scope of the presentinvention.

FIG. 13 illustrates an exemplary three-phase power supply circuit 1300including an output filter capacitor. This exemplary three-phase powersupply circuit 1300 is identical to the three-phase power supply circuit700, however bridge rectifier 702 has been replaced by synchronousbridge rectifier 1310 which is illustrated in FIG. 14 and described indetail below.

FIG. 14 illustrates an exemplary synchronous bridge rectifier 1310including six MOSFETs from FIG. 13 . In this example embodiment, therectifier diodes D7-D12 731-736 of bridge rectifier 702 of FIG. 7 havebeen replaced by MOSFETs Q10-Q15 1410-1415. Synchronous bridge rectifier1310 provides output VOUT 1404 with reference to ground GND 1405.

The start endings of the step-down transformers of FIG. 13 are connectedto the midpoint of each leg of the three-phase synchronous bridgerectifier 1310. OUT+ of phase 0 LLC voltage convertor 711 is coupled toϕA 1401, OUT+ of phase 1 LLC voltage convertor 712 is coupled to ΦB1402, and OUT+ of phase 2 LLC voltage convertor 713 is coupled to ΦC1403.

The phase A leg of three-phase synchronous bridge rectifier 1310includes Q10 1410 controlled by syncA 1430, syncA_U 1431, and R10 1420,and Q13 1413 controlled by syncA_N 1432 and R13 1423, with ΦA 1401provided between Q10 1410 and Q13 1413. The phase B leg of three-phasesynchronous bridge rectifier 1310 includes Q11 1411 controlled by syncB1433, syncB_U 1434, and R11 1421, and Q14 1414 controlled by syncB_N1435 and R14 1424, with ϕB 1402 provided between Q11 1411 and Q14 1414.The phase C leg of three-phase synchronous bridge rectifier 1310includes Q12 1412 controlled by syncC 1437, syncC_U 1438, and R12 1422,and Q15 1415 controlled by syncC_N 1439 and R15 1425, with ϕC 1403provided between Q12 1412 and Q15 1415.

Drive signals syncA 1430 and syncA—N 1432 should be in phase withcontrol signals AA0 760 and BBO 770 respectively. Drive signals syncB1433 and syncB N_1435 should be in phase with control signals AA1 761and BB1 771 respectively. Drive signals syncC 1436 and syncC_N 1438should be in phase with control signals AA2 762 and BBO 772respectively.

In order to avoid reverse current through MOSFETs Q10-Q15 1410-1415 fromdrain to source, the on time of the synchronous MOSFETs should be lessthan the resonant half time of the corresponding phase when theswitching frequency of the converter is less than or equal to theresonant frequency of the converter. When the switching frequency ishigher than the resonant frequency, the on time and the phasing arecarefully selected to avoid reverse current. Common commerciallyavailable circuits may be used to drive the synchronous rectifiers.

FIG. 15 illustrates an exemplary single-phase LLC resonant voltageconvertor 1500 for use within the three-phase power supply circuit 1300of FIG. 13 . This example single-phase LLC resonant voltage convertor1500 is identical to the single-phase LLC resonant voltage converter 600of FIG. 6 , however step-down transformer TX1 646 of FIG. 6 is providedexternal to LLC resonant voltage convertor 1500.

In this example embodiment, an input voltage is applied to inputs DC+1506 and DC-1508 of the voltage convertor 1500. In some exampleembodiments the input voltage may be provided by a power factorcorrection circuit.

Switch Q1 1541 and diode D1 1531 make up a first half-bridge and switchQ2 1542 and diode D2 1532 make up a second half-bridge. Diodes D1 131and D2 1532 are blocking diodes which block current when switches Q11541 and Q2 1542 are turned on simultaneously. Resistor R10 1557 toground is included between diodes D1 1531 and D2 1532.

Switch Q1 1541 is driven by isolated driver E1 1521 and resistors R11551 and R2 1552. Switch Q2 1542 is driven by isolated driver E3 1522and resistors R3 1553 and R4 1554. Isolated drivers E1 1521 and E2 1522are both driven by square wave AA 1502 which is generated by controlcircuit 800 illustrated in FIG. 8 and described in detail above.

The maximum voltage stress on switches Q1 1541 and Q2 1542 is equal tohalf of the input voltage between DC+ 1506 and DC− 1508, while switch Q31543 experiences the entire voltage stress of the input voltage betweenDC+ 1506 and DC− 1508. In an example embodiment, when the input voltagebetween DC+ 1506 and DC− 1508 is 440 volts, switches Q1 1541 and Q2 1542may be rated for 300-400 volts, while Q3 1543 is rated for 600-650volts.

Switch Q3 1543 is configured to short diodes D1 1531 and D2 1532 when itis activated by isolated driver E3 1523 and resistors R5 1555 and R61556. Isolated driver E3 1523 is driven by square wave BB 1504 which isgenerated by control circuit 800 illustrated in FIG. 8 and described indetail above.

Each half-bridge drives one node of the primary P1 coil of an externalstep-down transformer through a capacitor/inductor pair. The firsthalf-bridge comprising switch Q1 1541 and diode D1 1531 drives a firstnode of the primary P1 coil of the external step-down transformerthrough split resonant components capacitor C1 1561 and inductor L11544, electrically coupled in series. The second half-bridge comprisingswitch Q2 1542 and diode D2 1532 drives a second node of the primary P1coil of the external step-down transformer through split resonantcomponents capacitor C2 1562 and inductor L2 1545, electrically coupledin series.

Output voltages OUT+ 1510 and OUT− 1512 are provided to first and secondnodes of the primary coil P1 of an external step-down transformer asdescribed above.

In other example embodiments, a single resonant tank comprising C2 1562and L2 1545 may be used, in which case the value of C2 1562 will be halfof the value required by the split arrangement illustrated in FIG. 15 ,and the value of L2 1545 will be double the value required by the splitarrangement illustrated in FIG. 15 .

FIG. 16 illustrates an exemplary three-phase power supply circuit 1600with step-down transformers external to the three single-phase LLCresonant voltage convertors 1611, 1612, and 1613. This examplethree-phase power supply circuit 1600 is identical to that illustratedin FIG. 13 , however here the three step-down transformers T1 1631, T21632, and T3 1633 are provided external to the LLC voltage convertors1611, 1612, and 1613 which are illustrated by FIG. 15 and described indetail above.

In some example embodiments, step-down transformers T1 1631, T2 1632,and T3 1633 may utilize a unified core body. This embodiment isillustrated in FIGS. 17A, 17B, 18A, and 18B and described in detailbelow.

Bridge rectifier 1614 may comprise a diode bridge rectifier, such asbridge rectifier 702 of FIG. 7 , synchronous bridge rectifier 1310 ofFIG. 14 , or the like all within the scope of the present invention.

FIGS. 17A and 17B illustrate an exemplary unified core body 1700 for athree-phase power supply circuit such as that of FIG. 16 . In theseexample embodiments, a unified core body 1700 is configured to supportthree inductors and three transformers which are formed by a pluralityof windings. Unified core body 1700 includes air gaps 1710 whichinfluence various parameters of the inductors and transformers supportedby the core body 1700. The air gaps 1710 are provided to storemagnetizing energy in order to achieve zero-voltage switching (ZVS) forthe active switches within the power supply. Unified core body 1700 alsoincludes inductor return leg 1720 and transformer return leg 1730. Inthis embodiment, the return legs do not require air gaps.

Inductor return leg 1720 provides a return path for magnetic flux fromthe three inductors. Transformer return leg 1730 provides a return pathfor magnetic flux from the three transformers.

In an example embodiment, unified core body 1700 has a plurality of corelegs (here three are illustrated). Each core leg has a first and secondend, which each extend in a direction of central axes of the pluralityof windings and around which the plurality of windings are wound suchthat magnetic fluxes are produced in the plurality of core legs whencurrent flows through the plurality of windings.

FIGS. 18A and 18B illustrate an exemplary three-phase magnetics assembly1800 including three step-down transformers and three inductors for thethree-phase power supply circuit 1600 of FIG. 16 .

In this example embodiment unified core body 1810 has been populatedwith three transformers 1812, 1814, and 1816, along with three inductors1802, 1804, and 1806. FIG. 18B also illustrates inductor return leg 1822and transformer return leg 1820.

Since the inductors and transformers support large currents, eachcontributes to some amount of core loss from the magnetic flux withintheir cores. In order to minimize this core loss all three inductors andthree transformers are integrated together into three-phase magneticsassembly 1800. A common inductor return leg is provided for the threeinductors, and a common transformer return leg 1820 is provided for thethree transformers 1812, 1814, and 1816, as illustrated in FIGS. 17A and17B. Magnetic flux from the three phases within the single return legs1820 acts to cancel each other out since the phases are separated by120-degrees, thus reducing core loss within three-phase magneticsassembly 1800.

FIG. 19 illustrates exemplary magnetic fluxes within each of thetransformers and inductors and a common return leg within an exemplarypower supply circuit incorporating the three-phase magnetics assembly ofFIGS. 18A and 18B.

In an example embodiment, magnetic flux from each of the three inductorsis sinusoidal and offset by 120-degrees, so that the combined magneticflux in the return leg from the three inductors cancels itself out toessentially zero in ideal conditions. The magnetic flux in eachtransformer winding is triangular and offset by 120-degrees, so that thecombined magnetic flux from the three transformer phases act topartially cancel each other out, and reduce the magnetic flux within thetransformer return leg to ⅓ that of the flux in each individualtransformer leg.

FIG. 19 illustrates the relationship between inductor flux andtransformer flux within an exemplary power supply circuit incorporatinga three-phase magnetics assembly. In an example embodiment, such as thatillustrated in FIGS. 18A and 18B, magnetic flux within the threeinductors 1802, 1804, and 1806 has a sinusoidal shape, while magneticflux within the three transformers 1812, 1814, and 1816 has a triangularshape. Each phase is offset by 120 degrees or 2π/3.

Graph 1910 illustrates current within the three inductors 1802, 1804,and 1806. These current waveforms 1911, 1912, and 1913 are sinusoidal inshape and offset by 120 degrees or 2π/3. Graph 1920, comprisingwaveforms 1921, 1922, and 1923 illustrate the currents through theoutput rectifier diodes. Graph 1930 illustrates the output voltage VOUT1931.

Graph 1940 illustrates magnetic fluxes within each of the threetransformers and graph 1950 illustrates the magnetic flux within thecommon return leg within an exemplary power supply circuit incorporatinga three-phase magnetics assembly. As discussed above, each transformerhas a magnetic flux with a triangular waveform offset from each other by120 degrees or 2π/3. Here the magnetic flux within first phasetransformer T1 1631 is illustrated by waveform 1941, the magnetic fluxwithin second phase transformer T2 1632 is illustrated by waveform 1942,and the magnetic flux within third phase transformer T3 1633 isillustrated by waveform 1943. The magnetic flux through transformerreturn leg 1820 is illustrated by waveform 1951.

When these three magnetic fluxes are combined within common return leg1820, the amplitude of the combined fluxes is ⅓ that of each individualtransformer with a frequency three time that of the individualtransformers. By combining the magnetic fluxes from the threetransformers into a single transformer return leg, the amplitude of theflux is reduced by 2/3 and directly reduces core losses in the assembly.

FIG. 20 illustrates a portion of a three-phase power supply circuit 2000including an output inductor 2030. This example embodiment may beapplied to any of the three-phase power supply circuits describedherein.

In this example, bridge rectifier 2010 drives output voltage VOUT 2020across output filter capacitor C10 2040 driving load resistance RLOAD2050 through output inductor 2030 which is electrically coupled betweenthe output of bridge rectifier 2010 and output filter capacitor 2040.The addition of output inductor 2030 ensures current balance between thephases of three-phase power supply circuit 2000 in the situation whereresonant parameters are mismatched.

The included descriptions and figures depict specific embodiments toteach those skilled in the art how to make and use the best mode. Forthe purpose of teaching inventive principles, some conventional aspectshave been simplified or omitted. Those skilled in the art willappreciate variations from these embodiments that fall within the scopeof the invention. Those skilled in the art will also appreciate that thefeatures described above may be combined in various ways to formmultiple embodiments. As a result, the invention is not limited to thespecific embodiments described above, but only by the claims and theirequivalents.

1. A three-phase power supply circuit comprising: three LLC resonantvoltage convertors; three step-down transformers, each having a primaryand secondary coil, and each primary and secondary coil having a firstnode and a second node, wherein each step-down transformer iselectrically coupled with one of the three LLC resonant voltageconvertors by the first and second nodes of the primary coils; and abridge rectifier electrically coupled with the first node of thesecondary coil of each of the three step-down transformers; wherein thesecond nodes of the secondary coils of each of the three step-downtransformers are electrically coupled together.